FEATURES
Selectable 2-, 3-, 4-, or 5-phase operation at up to 1 MHz per phase
±7.7 mV worst-case differential sensing error over temperature
Logic-level PWM outputs for interface to external high-power drivers
Active current balancing between all output phases
Built-in power good/crowbar blanking supports on-the-fly VID code changes
Digitally programmable 0.5 V to 1.6 V output— supports both VR10.x and VR11 specifications
Programmable short-circuit protection with programmable latch-off delay
APPLICATIONS
Desktop PC power supplies for
Next generation Intel® processors VRM modules
GENERAL DESCRIPTION
The ADP31891 is a highly efficient multi-phase synchronous buck switching regulator controller optimized for converting a 12 V main supply into the core supply voltage required by high performance Intel processors. It uses an internal 8-bit DAC to read a voltage identification (VID) code directly from the
processor, which is used to set the output voltage between 0.5 V and 1.6 V.
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
Synchronous Buck Controller
ADP3189
This device uses a multi-mode PWM architecture to drive the logic-level outputs at a programmable switching frequency that can be optimized for VR size and efficiency. The phase relation-ship of the output signals can be programmed to provide 2-, 3-, 4-, or 5-phase operation, allowing for the construction of up to five complementary buck switching stages.
The ADP3189 also includes programmable no-load offset and slope functions to adjust the output voltage as a function of the load current, so it is optimally positioned for a system transient. The ADP3189 also provides accurate and reliable short-circuit protection, adjustable current limiting, and a delayed power good output that accommodates on-the-fly output voltage changes requested by the CPU.
ADP3189 is specified over the extended commercial tem-perature range of 0°C to +85°C and is available in a 40-lead LFCSP package.
1
Protected by U.S. Patent Number 6,683,441; others pending.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com Fax: 781.461.3113 ©2005 Analog Devices, Inc. All rights reserved.
ADP3189
TABLE OF CONTENTS
Functional Block Diagram..............................................................3 Specifications.....................................................................................4 Test Circuits.......................................................................................7 Absolute Maximum Ratings............................................................8 ESD Caution..................................................................................8 Pin Configuration and Function DescriptionS............................9 Typical Performance Characteristics...........................................11 Theory of Operation......................................................................12 Start-Up Sequence......................................................................12 Phase Detection Sequence.........................................................12 Master Clock Frequency............................................................13 Output Voltage Differential Sensing........................................13 Output Current Sensing............................................................13 Active Impedance Control Mode.............................................13 Current Control Mode and Thermal Balance........................13 Voltage Control Mode................................................................14 Delay Timer.................................................................................14 Soft Start......................................................................................14 Current Limit, Short Circuit, and Latch-Off Protection.......15 Dynamic VID..............................................................................15 Power Good Monitoring...........................................................15 Output Crowbar.........................................................................16 Output Enable and UVLO........................................................16 Thermal Monitoring..................................................................16 Application Information................................................................22 Setting the Clock Frequency.....................................................22 Soft Start Delay Time.................................................................22
Current Limit Latch-Off Delay Times.....................................22 Inductor Selection......................................................................23 Designing an Inductor...............................................................23 Selecting a Standard Inductor..............................................23 Current Sense Amplifier............................................................24 Inductor DCR Temperature Correction.................................24 Load Line Setting........................................................................25 Output Offset..............................................................................26 COUT Selection.............................................................................26 Power MOSFETs.........................................................................27 Ramp Resistor Selection............................................................28 COMP Pin Ramp.......................................................................28 Current Limit SetPoint..............................................................29 Feedback Loop Compensation Design....................................29 CIN Selection and Input Current di/dt Reduction..................31 Thermal Monitor Design..........................................................31 Tuning the ADP3189.................................................................32 DC Loadline Setting..............................................................32 AC Loadline Setting...............................................................33 Initial Transient Setting.........................................................33 Layout and Component Placement.........................................34 General Recommendations..................................................34 Power Circuitry Recommendations....................................34 Signal Circuitry Recommendations....................................34 Outline Dimensions.......................................................................35 Ordering Guide..........................................................................35
REVISION HISTORY
7/05—Revision 0: Initial Version
Rev. 0 | Page 2 of 36
ADP3189
VCCRT12FUNCTIONAL BLOCK DIAGRAM
RAMPADJ13ADP318931UVLOSHUTDOWNAND BIASOSCILLATORSETEN19ODGND18CMPRESET30PWM1850mVEN1DAC + 150mVCSREFCMPCURRENTBALANCINGCIRCUITCMPRESET2-/3-/4-/5-PHASEDRIVER LOGICRESET29PWM228PWM3CMPDAC– 250mVCMPPWRGD2DELAYRESET27PWM4RESETCURRENTLIMIT26PWM5CROWBAR25SW1TTSENSE10VRHOT9VRFAN8THERMALTHROTTLINGCONTROL24SW223SW322SW421SW517CSCOMPILIMIT11DELAY7CURRENTLIMITCIRCUIT15CSREF16CSSUMCOMP5+4FBPRECISIONREFERENCEFBRTN314LLSET–BOOTVOLTAGE& SOFT-STARTCONTROL6SSVIDSEL40323334VIDDAC05626-0013536373839VID7VID6VID5VID4VID3VID2VID1VID0
Figure 1.
Rev. 0 | Page 3 of 36
ADP3189
SPECIFICATIONS
VCC = 12 V, FBRTN = GND, TA = 0°C to 85°C, unless otherwise noted.1Table 1.
Parameter Symbol Conditions ERROR IAMPLIFER Output Voltage Range2VCOMP Accuracy VFB Relative to nominal DAC output, referenced to
FBRTN,
LLSET = CSREF, Figure 2,
VIDSEL = GND, VIDSEL = 1.25 V,
VID Range 1.00625 V to 1.60000 V
VFB(BOOT) In start-up Load Line Positioning Accuracy CSREF − LLSET = 80 mV Differential Non-Linearity
VCC = 10 V to 14 V Line Regulation ΔVFB
Input Bias Current IFB FBRTN Current IFBRTN Output Current ICOMP FB forced to VOUT − 3% Gain Bandwidth Product GBW(ERR) COMP = FB Slew Rate COMP = FB LLSET Input Voltage Range VLLSET Relative to CSREF LLSET Input Bias Current ILLSET BOOT Voltage Hold Time tBOOT CDELAY = 10 nF
IIVD NPUTS
Input Low Voltage VIL(VID) VIIDx, VDSEL Input High Voltage VIH(VID) VIDx, IVDSEL
2
Max VIH for VID on Fly Input Current IIN(VID) VID Transition Delay Time2 VID code change to FB change
VID code change to PWM going low No CPU Detection Turn-Off Delay
2
Time
IOSCLLATOR
2
Frequency RangefOSC Frequency Variation fPHASE TA = 25°C, RT = 243 kΩ, 5-phase TA = 25°C, RT = 113 kΩ, 5-phase TA = 25°C, RT = 51 kΩ, 5-phase Output Voltage VRT RT = 243 kΩ to GND RAMPADJ Output Voltage VRAMPADJ RAMPADJ − FB RAMPADJ Input Current Range IRAMPADJ CURRENT SENSE AMPLIFIER Offset Voltage VOS(CSA) CSSUM − CSREF, Figure 3Input Bias Current IBIAS(CSSUM) Gain Bandwidth Product GBW(CSA) CSSUM = CSCOMP Slew Rate CCSCOMP = 10 pF Input Common-Mode Range CSSUM and CSREF Output Voltage Range Output Current ICSCOMP Current Limit Latch-Off Delay Time tOC(DELAY) CDELAY = 10 nF
Rev. 0 | Page 4 of 36
Min Typ Max Unit
0.95 3.95 V
−7.7 −7.7 +7.7 mV +7.7 mV 1.092 1.1 1.108 V −78 −80 −82 mV
−1 +1 LSB 0.003 % 13.5 15 16.5 μA 125 200 μA 500 μA 20 MHz 25 V/μs −250 +250 mV −120 +120 nA 2 ms
0.4 V 0.8 V 1.26 V −1 μA 200 ns 200 ns 0.25 5 MHz 180 200 220 kHz 400 kHz 800 kHz 1.6 1.7 1.8 V −50 +50 mV 1 50 μA −1.0 +1.0 mV −50 +50 nA 10 MHz 10 V/μs 0 3 V 0.05 2.8 V 500 μA 8 ms
ADP3189
Parameter Symbol Conditions CURRENT BALANCE AMPLIFIER Common Mode Range VSW(X)CM Input Resistance RSW(X) SWx = 0 V Input Current ISW(X) SWx = 0 V Input Current Matching SWx = 0 V ΔISW(X) CURRENT IILMT COMPARATOR Output Voltage VILIMIT RILIMIT = 143 kΩ Output Current IILIMIT RILIMIT = 143 kΩ 2Maximum Output Current Current Limit Threshold Voltage VCL VCSREF − VCSCOMP, RILIMIT = 143 kΩ Current Limit Setting Ratio VCL/IILIMIT DELAY ITMER Normal Mode Output Current IDELAY Output Current in Current Limit IDELAY(CL) Threshold Voltage VDELAY(TH) SOFT START Output Current ISS During start-up ENABLE INPUT Threshold Voltage VTH(EN) Hysteresis VHYS(EN) Input Current IIN(EN) Delay Time tDELAY(EN) EN > 950 mV, CDELAY = 10 nF OD OUTPUT Output Low Voltage VOL(OD) Output High Voltage VOH(OD) Internally limited I VRFAN (SINK) = −4 mA I VRHOT (SINK) = −4 mA Relative to nominal DAC output Relative to nominal DAC output IPWRGD(SINK) = −4 mA CDELAY = 10 nF Relative to nominal DAC output Relative to FBRTN Overvoltage to PWM going low IPWM(SINK) = −400 μA IPWM(SOURCE) = 400 μA
Rev. 0 | Page 5 of 36
Min Typ Max Unit −600 +200 mV 35 50 65 kΩ 2.5 4.0 5.5 μA −5 +5 % 1.6 1.7 1.8 V 12 μA 60 μA 105 120 135 mV 10 mV/μA 12 15 18 μA 3.0 3.75 4.5 μA 1.6 1.7 1.8 V 12 15 18 μA 800 850 900 mV 80 100 120 mV −1 +1 μA 2 ms 100 500 mV V 4 5 THERMAL THROTTLING CONTROL TTSENSE Voltage Range TTSENSE VRFAN Threshold Voltage TTSENSE VRHOT Threshold Voltage TTSENSE Hysteresis TTSENSE Input Current VRFAN Output Low Voltage VOL(VRFAN) VRHOT Output Low Voltage VOL(VRHOT) POWER GOOD COMPARATOR Undervoltage Threshold VPWRGD(UV) Overvoltage Threshold VPWRGD(OV) Output Low Voltage VOL(PWRGD) Power Good Delay Time 2 During Soft Start VID Code Changing VID Code Static Crowbar Trip Point VCROWBAR Crowbar Reset Point Crowbar Delay Time tCROWBAR VID Code Changing VID Code Static PWM OUTPUTS Output Low Voltage VOL(PWM) Output High Voltage VOH(PWM) 0 5.3 V 1.08 1.11 1.14 V 780 810 840 mV 55 mV −105 −120 −135 μA 150 300 mV 150 300 mV −200 −250 −300 mV 100 150 200 mV 150 300 mV 2 ms 100 400 μs 200 ns 100 150 200 mV 320 375 430 mV 100 400 μs 400 ns 160 500 mV 4.0 5 V
ADP3189
Parameter Symbol Conditions SUPPLY DC Supply Current UVLO Threshold Voltage VUVLO VCC rising UVLO Hysteresis
Min Typ Max Unit
6 10 mA 7 7.4 7.8 V 0.4 0.6 0.8 V 12
All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). Guaranteed by design or bench characterization, not tested in production.
Rev. 0 | Page 6 of 36
ADP3189
8-BIT CODE+12V1μF100nFTEST CIRCUITS
40L01234567EDDDDDDDDCSIIIIIIIIC1DVVVVVVVVVI1.25VENVPWM1PWRGDPWM2FBRTNPWM3FBPWM41kΩCOMPPWM5SSADP3189SW110nFDELAYSW210nFVRFANSW3VRHOTJSW4TTSENSEDPAFMMSW5TPTIEUOMMESRSCDILTALSSSNDCIRRLCCCGON250kΩ20kΩ100nFFigure 2. Closed-Loop Output Voltage Accuracy
ADP3189VCC12V31CSCOMP1739kΩ100nFCSSUM161kΩCSREF151.25VGNDVOS =CSCOMP– 1.25V4030180-62650Figure 3. Current Sense Amplifier VOS
200-62650
Rev. 0 | Page 7 of 36
ADP3189VCC12V31COMP510kΩFB4LLSET14–ΔVCSREF15VID+DAC1.25VGND40180-62650 Figure 4. Positioning Voltage
ADP3189
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Rating may cause permanent damage to the device. This is a stress VCC −0.3 V to +15 V rating only and functional operation of the device at these or FBRTN −0.3 V to +0.3 V any other conditions above those indicated in the operational PWM3 to PWM5, RAMPADJ −0.3 V to VCC + 0.3 V section of this specification is not implied. Exposure to absolute SW1 to SW5 −5 V to +25 V maximum rating conditions for extended periods may affect <200 ns −10 V to +25 V device reliability. Absolute maximum ratings apply individually All Other Inputs and Outputs −0.3 V to +5.5 V only, not in combination. Unless otherwise specified all other Stresses above those listed under Absolute Maximum Ratings
Storage Temperature −65°C to +150°C voltages re referenced to GND. Operating Ambient Temperature Range 0°C to +85°C
Operating Junction Temperature 125°C
Thermal Impedance (θJA)
100°C/W Lead Temperature
Soldering (10 sec) 300°C Infrared (15 sec) 260°C
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. 0 | Page 8 of 36
ADP3189
VIDSELVID0VID1VID2VID3VID4VID5VID6VID7VCC40393837363534333231PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
ENPWGRDFBRTNFBCOMPSSDELAYVRFANVRHOTTTSENSE1PIN 12INDICATOR34ADP31895TOP VIEW6(Not to Scale)789101112131415161718192030292827262524232221PWM1PWM2PWM3PWM4PWM5SW1SW2SW3SW4SW5ILIMITRTRAMPADJLLSETCSREFCSSUMCSCOMPGNDODNC05626-005
Figure 5. Pin Configuration
Table 3. Pin Function Descriptions Pin No. 1 2 3 4 Mnemonic EN PWRGD FBRTN FB Description Power Supply Enable Input. Pulling this pin to GND disables the PWM outputs and pulls the PWRGD output low. Power Good Output. Open-drain output that signals when the output voltage is outside of the proper operating range. Feedback Return. VID DAC and error amplifier reference for remote sensing of the output voltage. Feedback Input. Error amplifier input for remote sensing of the output voltage. An external resistor between this pin and the output voltage sets the no-load offset point. Error Amplifier Output and Compensation Point. Soft Start Delay Setting Input. An external capacitor connected between this pin and GND sets the soft start ramp-up time. Delay Timer Setting Input. An external capacitor connected between this pin and GND sets the overcurrent latch-off delay time, BOOT voltage hold time, EN delay time, and PWRGD delay time. VR Fan Activation Output. Active high open drain output that signals when the temperature at the monitoring point connected to TTSENSE exceeds the programmed VRFAN temperature threshold. VR Hot Output. Active high open drain output that signals when the temperature at the monitoring point connected to TTSENSE exceeds the programmed VRHOT temperature threshold. VR Hot Thermal Throttling Sense Input. An NTC thermistor between this pin and GND is used to remotely sense the temperature at the desired thermal monitoring point. Current Limit Set Point. An external resistor from this pin to GND sets the current limit threshold of the converter. Frequency Setting Resistor Input. An external resistor connected between this pin and GND sets the oscillator frequency of the device. PWM Ramp Current Input. An external resistor from the converter input voltage to this pin sets the internal PWM ramp. Output Load Line Programming Input. This pin can be directly connected to CSCOMP, or it can be connected to the center point of a resistor divider between CSCOMP and CSREF. Connecting LLSET to CSREF disables positioning. Current Sense Reference Voltage Input. The voltage on this pin is used as the reference for the current sense amplifier and the power good and crowbar functions. This pin should be connected to the common point of the output inductors. Current Sense Summing Node. External resistors from each switch node to this pin sum the average inductor currents together to measure the total output current. Current Sense Compensation Point. A resistor and capacitor from this pin to CSSUM determines the gain of the current sense amplifier and the positioning loop response time. Ground. All internal biasing and the logic output signals of the device are referenced to this ground. Output Disable Logic Output. This pin is actively pulled low when the ADP3189 EN input is low or when VCC is below its UVLO threshold to signal to the Driver IC that the driver high-side and low-side outputs should go low. No Connect. Rev. 0 | Page 9 of 36
5 COMP 6 SS 7 8 9 10 11 12 13 14 15 16 17 18 19 DELAY VRFAN VRHOT TTSENSE ILIMIT RT RAMPADJ LLSET CSREF CSSUM CSCOMP GND OD20 NC ADP3189
Pin No. 21 to 25 26 to 30 31
32 to 39
Mnemonic SW5 to SW1 PWM5 to PMW1 VCC
VID7 to VID0
40 VIDSEL
Description
Current Balance Inputs. Inputs for measuring the current level in each phase. The SW pins of unused phases should be left open.
Logic-Level PWM Outputs. Each output is connected to the input of an external MOSFET driver such as the ADP3120. Connecting the PWM3, PWM4, and/or PWM5 outputs to VCC will cause that phase to turn off, allowing the ADP3189 to operate as a 2-, 3-, 4-, or 5-phase controller. Supply Voltage for the Device.
Voltage Identification DAC Inputs. These eight pins are pulled down to GND, providing a logic zero if left open. When in normal operation mode, the DAC output programs the FB regulation voltage from 0.5 V to 1.6 V (see Table 4).
VID DAC Selection Pin. The logic state of this pin determines whether the internal VID DAC decodes VID0 to VID7 as extended VR10 or VR11 inputs.
Rev. 0 | Page 10 of 36
ADP3189
5.0k4.5kOSCILATOR FREQUENCY (kHz)TYPICAL PERFORMANCE CHARACTERISTICS
5.0k4.5k4.0k3.5k3.0k2.5k2.0k1.5k1.0k05626-006OSCILATOR FREQUENCY (kHz)4.0k3.5k3.0k2.5k2.0k1.5k1.0k0.5k05.86.06.2SUPPLY CURRENT (μA)6.405626-0070.5k00200500Rt (kΩ)80010006.6
Figure 6. Master Clock Frequency vs. RT Figure 7. Oscillator Frequency vs .Supply Current
Rev. 0 | Page 11 of 36
ADP3189
THEORY OF OPERATION
The ADP3189 combines a multimode, fixed frequency PWM control with multiphase logic outputs for use in 2-, 3-, 4-, and 5-phase synchronous buck CPU core supply power converters. The internal VID DAC is designed to interface with the Intel 8-bit VRD/VRM 11- and 7-bit VRD/VRM 10x-compatible CPUs. Multiphase operation is important for producing the high currents and low voltages demanded by today’s microproc-essors. Handling the high currents in a single-phase converter places high thermal demands on the components in the system, such as the inductors and MOSFETs.
The multimode control of the ADP3189 ensures a stable, high performance topology for the following: • • • • • • • • •
Balancing currents and thermals between phases High speed response at the lowest possible switching frequency and output decoupling
Minimizing thermal switching losses by using lower frequency operation
Tight load line regulation and accuracy
High current output from having up to 5-phase operation Reduced output ripple due to multiphase cancellation PC board layout noise immunity
Ease of use and design due to independent component selection
Flexibility in operation for tailoring design to low cost or high performance
VR READY(ADP3189 PWRGD)50μsCPUVID INPUTSVID INVALIDTD5VID VALID05626-00812VSUPPLYUVLOTHRESHOLDVTT I/O(ADP3189 EN)0.85VDELAYVDELAY(TH)(1.7V)SS1.0VVBOOT(1.1V)TD3VVIDVCC_CORETD1VBOOT(1.1V)VVIDTD4TD2
Figure 8. System Start-Up Sequence
PHASE DETECTION SEQUENCE
During start-up, the number of operational phases and their phase relationship is determined by the internal circuitry moni-toring the PWM outputs. Normally, the ADP3189 operates as a 5-phase PWM controller. Connecting the PWM5 pin to VCC programs a 4-phase operation, and connecting the PWM5 pin and PWM4 pin to VCC programs a 3-phase operation. For 2-phase operation, connect PWM5, PWM4, and PWM3 to VCC.
Prior to soft start, while EN is low, the PWM3, PWM4, and PWM5 pins sink approximately 100 μA. An internal compara-tor checks each pin’s voltage vs. a threshold of 3.15 V. If the pin is tied to VCC, it is above the threshold. Otherwise, an internal current sink pulls the pin to GND, which is below the threshold. PWM1 and PWM2 are low during the phase detection interval, which occurs during the first five clock cycles of the internal oscillator. After this time, if the remaining PWM outputs are not pulled to VCC, the 100 μA current sink is removed, and they function as normal PWM outputs. If they are pulled to VCC, the 100 μA current source is removed, and the outputs are put into a high-impedance state.
The PWM outputs are logic-level devices intended for driving external gate drivers such as the ADP3120. Since each phase is monitored independently, operation approaching 100% duty cycle is possible. Also, more than one output can be on at the same time to allow overlapping phases.
START-UP SEQUENCE
The ADP3189 follows the VR11 start-up sequence shown in Figure 8. After both the EN and UVLO conditions are met, the DELAY pin goes through one cycle (TD1). After this cycle, the internal oscillator is enabled. The first five clock cycles are blanked from the PWM outputs and used for phase detection as explained in the Phase Detection Sequence section. Then, the soft start ramp is enabled (TD2), and the output comes up to the boot voltage of 1.1 V. The boot hold time is determined by the DELAY pin as it goes through a second cycle (TD3). During TD3, the processor VID pins settle to the required VID code. When TD3 is over, the ADP3189 soft starts either up or down to the final VID voltage (TD4). After TD4 has been completed and the PWRGD masking time (equal to VID on the fly masking) is finished, a third ramp on the DELAY pin sets the PWRGD blanking (TD5).
Rev. 0 | Page 12 of 36
ADP3189
An additional resistor divider connected between CSREF and CSCOMP, with the mid point connected to LLSET, can be used to set the load line required by the microprocessor. The current information is then given as CSREF – LLSET. This difference signal is used internally to offset the VID DAC for voltage positioning. The difference between CSREF and CSCOMP is then used as a differential input for the current-limit comparator. This allows for the load line to be set independ-ently of the current-limit threshold. In the event that the current limit threshold and load line are not independent, the resistor divider between CSREF and CSCOMP can be removed and the CSCOMP pin can be directly connected to LLSET. To disable voltage positioning entirely (that is, no load line) connect LLSET to CSREF.
To provide the best accuracy for sensing current, the CSA is designed to have a low offset input voltage. Also, the sensing gain is determined by external resistors, so that it can be made extremely accurate.
MASTER CLOCK FREQUENCY
The clock frequency of the ADP3189 is set with an external resistor connected from the RT pin to ground. The frequency follows the graph in Figure 6. To determine the frequency per phase, the clock is divided by the number of phases in use. If all phases are in use, divide by 5. If PWM5 is tied to VCC, then divide the master clock by 4 for the frequency of the remaining phases. If PWM4 and PWM5 are tied to VCC, then divide by 3. If PWM3, PWM4, and PWM5 are tied to VCC, then divide by 2.
OUTPUT VOLTAGE DIFFERENTIAL SENSING
The ADP3189 combines differential sensing with a high
accuracy VID DAC and reference and a low offset error ampli-fier. This maintains a worst-case specification of ±7.7 mV differential sensing error over its full operating output voltage and temperature range. The output voltage is sensed between the FB pin and FBRTN pin. FB should be connected through a resistor to the regulation point, usually the remote sense pin of the microprocessor. FBRTN should be connected directly to the remote sense ground point. The internal VID DAC and precision reference are referenced to FBRTN, which has a minimal current of 125 μA to allow accurate remote sensing. The internal error amplifier compares the output of the DAC to the FB pin to regulate the output voltage.
ACTIVE IMPEDANCE CONTROL MODE
For controlling the dynamic output voltage droop as a function of output current, a signal proportional to the total output current at the LLSET pin can be scaled to be equal to the droop imped-ance of the regulator times the output current. This droop voltage is then used to set the input control voltage to the system. The droop voltage is subtracted from the DAC reference input voltage directly to tell the error amplifier where the output voltage should be. This allows enhanced feed-forward response.
OUTPUT CURRENT SENSING
The ADP3189 provides a dedicated current-sense amplifier (CSA) to monitor the total output current for proper voltage positioning vs. load current and for current-limit detection. Sensing the load current at the output gives the total average current being delivered to the load, which is an inherently more accurate method than peak current detection or sampling the current across a sense element such as the low-side MOSFET. This amplifier can be configured several ways, depending on the objectives of the system, as follows: • • •
Output inductor DCR sensing without a thermistor for lowest cost.
Output inductor DCR sensing with a thermistor for
improved accuracy with tracking of inductor temperature. Sense resistors for highest accuracy measurements.
CURRENT CONTROL MODE AND THERMAL BALANCE
The ADP3189 has individual inputs (SW1 to SW5) for each phase, which are used for monitoring the current of each phase. This information is combined with an internal ramp to create a current balancing feedback system that has been optimized for initial current balance accuracy and dynamic thermal balancing during operation. This current balance information is independent of the average output current information used for positioning as described in the Output Current Sensing section.
The magnitude of the internal ramp can be set to optimize the transient response of the system. It also monitors the supply volt-age for feed-forward control for changes in the supply. A resistor connected from the power input voltage to the RAMPADJ pin determines the slope of the internal PWM ramp. External resistors can be placed in series with individual phases to create an intentional current imbalance if desired, such as when one phase has better cooling and can support higher currents. Resistors RSW1 through RSW5 (see the Typical
Application Circuit in Figure 11) can be used for adjusting thermal balance. It is best to have the ability to add these resistors during the initial design, so ensure that placeholders are provided in the layout.
The positive input of the CSA is connected to the CSREF pin, which is connected to the output voltage. The inputs to the amplifier are summed together through resistors from the sensing element, such as the switch node side of the output inductors, to the inverting input, CSSUM. The feedback resistor between CSCOMP and CSSUM sets the gain of the amplifier, and a filter capacitor is placed in parallel with this resistor. The gain of the amplifier is programmable by adjusting the feedback resistor.
Rev. 0 | Page 13 of 36
ADP3189
To increase the current in any given phase, enlarge RSW for that phase (make RSW = 0 for the hottest phase and do not change during balancing). Increasing RSW to only 500 Ω makes a substantial increase in phase current. Increase each RSW value by small amounts to achieve balance, starting with the coolest phase first.
VOLTAGE CONTROL MODE
A high gain-bandwidth voltage mode error amplifier is used for the voltage-mode control loop. The control input voltage to the positive input is set via the VID logic according to the voltages listed in Table 4.
This voltage is also offset by the droop voltage for active positioning of the output voltage as a function of current, commonly known as active voltage positioning. The output of the amplifier is the COMP pin, which sets the termination voltage for the internal PWM ramps.
The negative input (FB) is tied to the output sense location with a resistor RB and is used for sensing and controlling the output voltage at this point. A current source from the FB pin flowing through RB is used for setting the no-load offset voltage from the VID voltage. The no-load voltage is negative with respect to the VID DAC. The main loop compensation is incorporated into the feedback network between FB and COMP.
Once the SS voltage is within 100 mV of the boot voltage, the boot voltage delay time (TD3) is started. The end of the boot voltage delay time signals the beginning of the second soft start time (TD4). The SS voltage now changes from the boot voltage to the programmed VID DAC voltage (either higher or lower) using the SS amplifier with the limited output current of 15 μA. The voltage of the FB pin follows the ramping voltage of the SS pin, limiting the inrush current during the transition from the boot voltage to the final DAC voltage. The second soft start time depends on the boot voltage, the programmed VID DAC voltage, and CSS.
If either EN is taken low or VCC drops below UVLO, DELAY and SS are reset to ground to be ready for another soft start cycle. Figure 9 shows typical start-up waveforms for the ADP3189.
132DELAY TIMER
The delay times for the start-up timing sequence are set with a capacitor from the DELAY pin to ground. In UVLO, or when EN is logic low, the DELAY pin is held at ground. After the UVLO and EN signals are asserted, the first delay time (TD1 in Figure 8) is initiated. A 15 μA current flows out of the DELAY pin to charge CDLY. A comparator monitors the DELAY voltage with a threshold of 1.7 V. The delay time is therefore set by the 15 μA charging a capacitor from 0 V to 1.7 V. This DELAY pin is used for multiple delay timings (TD1, TD3, and TD5) during the start-up sequence. Also, DELAY is used for timing the current limit latch off, as explained in the Current Limit, Short Circuit, and Latch-Off Protection section.
4CH1 1.0VCH3 1.0VCH2 1.0VCH4 10.0VM2.00msT 22.0%A CH1 500V05626-009
Figure 9. Typical Start-up Waveforms Channel 1: CSREF, Channel 2: DELAY, Channel 3: SS, Channel 4: Phase 1 Switch Node
SOFT START
The Soft Start times for the output voltage are set with a capacitor from the SS pin to ground. After TD1 and the phase detection cycle have been completed, the SS time (TD2 in Figure 8) starts. The SS pin is disconnected from GND, and the capacitor is charged up to the 1.1 V boot voltage by the SS amplifier, which has a limited output current of 15 μA. The voltage at the FB pin follows the ramping voltage on the SS pin, limiting the inrush current during start-up. The soft start time depends on the value of the boot voltage and CSS.
Rev. 0 | Page 14 of 36
ADP3189
CURRENT LIMIT, SHORT CIRCUIT, AND LATCH-OFF PROTECTION
The ADP3189 compares a programmable current-limit set point to the voltage from the output of the current-sense amplifier. The level of current limit is set with the resistor from the ILIMIT pin to ground. During operation, the voltage on ILIMIT is 1.7 V. The current through the external resistor is internally scaled to give a current limit threshold of 10 mV/μA. If the difference in voltage between CSREF and CSCOMP rises above the current limit threshold, the internal current limit amplifier controls the internal COMP voltage to maintain the average output current at the limit.
If the limit is reached and TD5 has completed, a latch-off delay time starts, and the controller shuts down if the fault is not removed. The current limit delay time shares the DELAY pin timing capacitor with the start-up sequence timing. However, during current limit, the DELAY pin current is reduced to 3.75 μA. A comparator monitors the DELAY voltage and shuts off the controller when the voltage reaches 1.7 V. Therefore, the current limit latch-off delay time is set by the current of 3.75 μA, charging the delay capacitor from 0 V to 1.7 V. This delay is four times longer than the delay time during the start-up sequence.
The current limit delay time starts only after the TD5 has completed. If there is a current limit during start-up, the
ADP3189 goes through TD1 to TD5, and then starts the latch-off time. Because the controller continues to cycle the phases during the latch-off delay time, if the short is removed before the 1.7 V threshold is reached, the controller returns to normal operation, and the DELAY capacitor is reset to GND.
The latch-off function can be reset by either removing and reapplying the supply voltage to the ADP3189, or by toggling the EN pin low for a short time. To disable the short circuit latch-off function, an external resistor should be placed in parallel with CDLY. This prevents the DELAY capacitor from charging up to the 1.7 V threshold. The addition of this resistor will cause a slight increase in the delay times.
During start-up, when the output voltage is below 200 mV, a secondary current limit is active. This is necessary because the voltage swing of CSCOMP cannot go below ground. This secondary current limit controls the internal COMP voltage to the PWM comparators to 1.5 V. This limits the voltage drop across the low-side MOSFETs through the current balance circuitry. An inherent per-phase current limit protects individual phases if one or more phases stop functioning because of a faulty component. This limit is based on the maximum normal mode COMP voltage. Typical overcurrent latch-off waveforms are shown in Figure 10.
1324CH1 1.0VCH3 1.0VCH2 1.0VCH4 10.0VM2.00msT 22.0%A CH1 500V05626-010
Figure 10. Overcurrent Latch-Off Waveforms Channel 1: CSREF, Channel 2: DELAY,
Channel 3: COMP, Channel 4: Phase 1 Switch Node
DYNAMIC VID
The ADP3189 has the ability to dynamically change the VID inputs while the controller is running. This allows the output voltage to change while the supply is running and supplying current to the load. This is commonly referred to as VID on-the-fly (OTF). A VID OTF can occur under light or heavy load conditions. The processor signals the controller by changing the VID inputs in multiple steps from the start code to the finish code. This change can be positive or negative.
When a VID input changes state, the ADP3189 detects the change and ignores the DAC inputs for a minimum of 200 ns. This time prevents a false code due to logic skew while the eight VID inputs are changing. Additionally, the first VID change initiates the PWRGD and crowbar blanking functions for a minimum of 100 μs to prevent a false PWRGD or crowbar event. Each VID change resets the internal timer.
POWER GOOD MONITORING
The power good comparator monitors the output voltage via the CSREF pin. The PWRGD pin is an open-drain output whose high level, when connected to a pull-up resistor, indicates that the output voltage is within the nominal limits specified based on the VID voltage setting. PWRGD goes low if the output voltage is outside of this specified range, if the VID DAC inputs are in no CPU mode, or whenever the EN pin is pulled low. PWRGD is blanked during a VID OTF event for a period of 400 μs to prevent false signals during the time the output is changing.
Rev. 0 | Page 15 of 36
ADP3189
The PWRGD circuitry also incorporates an initial turn-on delay time (TD5), based on the DELAY timer. Prior to the SS voltage reaching the programmed VID DAC voltage and the PWRGD masking time finishing, the PWRGD pin is held low. Once the SS pin is within 100 mV of the programmed DAC voltage, the capacitor on the DELAY pin begins to charge up. A comparator monitors the DELAY voltage and enables PWRGD when the voltage reaches 1.7 V. The PWRGD delay time is, therefore, set by a current of 15 μA, charging a capacitor from 0 V to 1.7 V.
THERMAL MONITORING
The ADP3189 includes a thermal monitoring circuit to detect when a point on the VR has exceeded two different user-defined temperatures. The thermal monitoring circuit requires an NTC thermistor to be placed between TTSENSE and GND. A fixed current of 120 μA is sourced out of the TTSENSE pin and into the thermistor. The current source is internally limited to 5 V. An internal circuit compares the TTSENSE voltage to a 1.11 V and a 0.81 V threshold, and outputs an open-drain signal at the VRFAN and VRHOT outputs, respectively. Once the voltage on the TTSENSE pin goes below its respective threshold, the open drain outputs assert high to signal the system that an overtem-perature event has occurred. Since the TTSENSE voltage changes slowly with respect to time, 55 mV of hysteresis is built into these comparators. The thermal monitoring circuitry does not depend on EN and is active when UVLO is above its threshold. When UVLO is below its threshold, VRFAN and VRHOT are forced low.
OUTPUT CROWBAR
As part of the protection for the load and output components of the supply, the PWM outputs are driven low, turning on the low-side MOSFETs, when the output voltage exceeds the upper crowbar threshold. This crowbar action stops once the output voltage falls below the release threshold of approximately 375 mV.
Turning on the low-side MOSFETs pulls down the output as the reverse current builds up in the inductors. If the output overvoltage is due to a short in the high-side MOSFET, this action current-limits the input supply or blows its fuse, protecting the microprocessor from being destroyed.
OUTPUT ENABLE AND UVLO
For the ADP3189 to begin switching, the input supply (VCC) to the controller must be higher than the UVLO threshold, and the EN pin must be higher than its 0.85 V threshold. This initiates a system start up sequence. If either UVLO or EN is less than their respective thresholds, the ADP3189 is disabled. This holds the PWM outputs at ground, shorts the DELAY capacitor to ground, and forces PWRGD and OD signals low. In the application circuit, the OD pin should be connected to the OD inputs of the ADP3120 driver. Grounding OD disables the drivers such that both DRVH and DRVL are grounded. This feature is important in preventing the discharge of the output capacitors when the controller is shut off. If the driver outputs were not disabled, a negative voltage can be generated during output due to the high current discharge of the output capacitors through the inductors.
Rev. 0 | Page 16 of 36
ADP3189
Table 4.VR11 and VR10.x VID Codes for the ADP3189
VR11 DAC CODES: VIDSEL = HIGH VR10.x DAC CODES: VIDSEL = LOW
OUTPUT VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6 OFF 0 0 0 0 0 0 0 0 N/A OFF 0 0 0 0 0 0 0 1 N/A 1.60000 0 0 0 0 0 0 1 0 0 1 0 1 0 1 1 1.59375 0 0 0 0 0 0 1 1 0 1 0 1 0 1 0 1.58750 0 0 0 0 0 1 0 0 0 1 0 1 1 0 1 1.58125 0 0 0 0 0 1 0 1 0 1 0 1 1 0 0 1.57500 0 0 0 0 0 1 1 0 0 1 0 1 1 1 1 1.56875 0 0 0 0 0 1 1 1 0 1 0 1 1 1 0 1.56250 0 0 0 0 1 0 0 0 0 1 1 0 0 0 1 1.55625 0 0 0 0 1 0 0 1 0 1 1 0 0 0 0 1.55000 0 0 0 0 1 0 1 0 0 1 1 0 0 1 1 1.54375 0 0 0 0 1 0 1 1 0 1 1 0 0 1 0 1.53750 0 0 0 0 1 1 0 0 0 1 1 0 1 0 1 1.53125 0 0 0 0 1 1 0 1 0 1 1 0 1 0 0 1.52500 0 0 0 0 1 1 1 0 0 1 1 0 1 1 1 1.51875 0 0 0 0 1 1 1 1 0 1 1 0 1 1 0 1.51250 0 0 0 1 0 0 0 0 0 1 1 1 0 0 1 1.50625 0 0 0 1 0 0 0 1 0 1 1 1 0 0 0 1.50000 0 0 0 1 0 0 1 0 0 1 1 1 0 1 1 1.49375 0 0 0 1 0 0 1 1 0 1 1 1 0 1 0 1.48750 0 0 0 1 0 1 0 0 0 1 1 1 1 0 1 1.48125 0 0 0 1 0 1 0 1 0 1 1 1 1 0 0 1.47500 0 0 0 1 0 1 1 0 0 1 1 1 1 1 1 1.46875 0 0 0 1 0 1 1 1 0 1 1 1 1 1 0 1.46250 0 0 0 1 1 0 0 0 1 0 0 0 0 0 1 1.45625 0 0 0 1 1 0 0 1 1 0 0 0 0 0 0 1.45000 0 0 0 1 1 0 1 0 1 0 0 0 0 1 1 1.44375 0 0 0 1 1 0 1 1 1 0 0 0 0 1 0 1.43750 0 0 0 1 1 1 0 0 1 0 0 0 1 0 1 1.43125 0 0 0 1 1 1 0 1 1 0 0 0 1 0 0 1.42500 0 0 0 1 1 1 1 0 1 0 0 0 1 1 1 1.41875 0 0 0 1 1 1 1 1 1 0 0 0 1 1 0 1.41250 0 0 1 0 0 0 0 0 1 0 0 1 0 0 1 1.40625 0 0 1 0 0 0 0 1 1 0 0 1 0 0 0 1.40000 0 0 1 0 0 0 1 0 1 0 0 1 0 1 1 1.39375 0 0 1 0 0 0 1 1 1 0 0 1 0 1 0 1.38750 0 0 1 0 0 1 0 0 1 0 0 1 1 0 1 1.38125 0 0 1 0 0 1 0 1 1 0 0 1 1 0 0 1.37500 0 0 1 0 0 1 1 0 1 0 0 1 1 1 1 1.36875 0 0 1 0 0 1 1 1 1 0 0 1 1 1 0 1.36250 0 0 1 0 1 0 0 0 1 0 1 0 0 0 1 1.35625 0 0 1 0 1 0 0 1 1 0 1 0 0 0 0 1.35000 0 0 1 0 1 0 1 0 1 0 1 0 0 1 1 1.34375 0 0 1 0 1 0 1 1 1 0 1 0 0 1 0 1.33750 0 0 1 0 1 1 0 0 1 0 1 0 1 0 1 1.33125 0 0 1 0 1 1 0 1 1 0 1 0 1 0 0 1.32500 0 0 1 0 1 1 1 0 1 0 1 0 1 1 1 1.31875 0 0 1 0 1 1 1 1 1 0 1 0 1 1 0 Rev. 0 | Page 17 of 36
ADP3189
VR11 DAC CODES: VIDSEL = HIGH VR10.x DAC CODES: VIDSEL = LOW
OUTPUT VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6 1.31250 0 0 1 1 0 0 0 0 1 0 1 1 0 0 1 1.30625 0 0 1 1 0 0 0 1 1 0 1 1 0 0 0 1.30000 0 0 1 1 0 0 1 0 1 0 1 1 0 1 1 1.29375 0 0 1 1 0 0 1 1 1 0 1 1 0 1 0 1.28750 0 0 1 1 0 1 0 0 1 0 1 1 1 0 1 1.28125 0 0 1 1 0 1 0 1 1 0 1 1 1 0 0 1.27500 0 0 1 1 0 1 1 0 1 0 1 1 1 1 1 1.26875 0 0 1 1 0 1 1 1 1 0 1 1 1 1 0 1.26250 0 0 1 1 1 0 0 0 1 1 0 0 0 0 1 1.25625 0 0 1 1 1 0 0 1 1 1 0 0 0 0 0 1.25000 0 0 1 1 1 0 1 0 1 1 0 0 0 1 1 1.24375 0 0 1 1 1 0 1 1 1 1 0 0 0 1 0 1.23750 0 0 1 1 1 1 0 0 1 1 0 0 1 0 1 1.23125 0 0 1 1 1 1 0 1 1 1 0 0 1 0 0 1.22500 0 0 1 1 1 1 1 0 1 1 0 0 1 1 1 1.21875 0 0 1 1 1 1 1 1 1 1 0 0 1 1 0 1.21250 0 1 0 0 0 0 0 0 1 1 0 1 0 0 1 1.20625 0 1 0 0 0 0 0 1 1 1 0 1 0 0 0 1.20000 0 1 0 0 0 0 1 0 1 1 0 1 0 1 1 1.19375 0 1 0 0 0 0 1 1 1 1 0 1 0 1 0 1.18750 0 1 0 0 0 1 0 0 1 1 0 1 1 0 1 1.18125 0 1 0 0 0 1 0 1 1 1 0 1 1 0 0 1.17500 0 1 0 0 0 1 1 0 1 1 0 1 1 1 1 1.16875 0 1 0 0 0 1 1 1 1 1 0 1 1 1 0 1.16250 0 1 0 0 1 0 0 0 1 1 1 0 0 0 1 1.15625 0 1 0 0 1 0 0 1 1 1 1 0 0 0 0 1.15000 0 1 0 0 1 0 1 0 1 1 1 0 0 1 1 1.14375 0 1 0 0 1 0 1 1 1 1 1 0 0 1 0 1.13750 0 1 0 0 1 1 0 0 1 1 1 0 1 0 1 1.13125 0 1 0 0 1 1 0 1 1 1 1 0 1 0 0 1.12500 0 1 0 0 1 1 1 0 1 1 1 0 1 1 1 1.11875 0 1 0 0 1 1 1 1 1 1 1 0 1 1 0 1.11250 0 1 0 1 0 0 0 0 1 1 1 1 0 0 1 1.10625 0 1 0 1 0 0 0 1 1 1 1 1 0 0 0 1.10000 0 1 0 1 0 0 1 0 1 1 1 1 0 1 1 1.09375 0 1 0 1 0 0 1 1 1 1 1 1 0 1 0 OFF N/A 1 1 1 1 1 0 1 OFF N/A 1 1 1 1 1 0 0 OFF N/A 1 1 1 1 1 1 1 OFF N/A 1 1 1 1 1 1 0 1.08750 0 1 0 1 0 1 0 0 0 0 0 0 0 0 1 1.08125 0 1 0 1 0 1 0 1 0 0 0 0 0 0 0 1.07500 0 1 0 1 0 1 1 0 0 0 0 0 0 1 1 1.06875 0 1 0 1 0 1 1 1 0 0 0 0 0 1 0 1.06250 0 1 0 1 1 0 0 0 0 0 0 0 1 0 1 1.05625 0 1 0 1 1 0 0 1 0 0 0 0 1 0 0 1.05000 0 1 0 1 1 0 1 0 0 0 0 0 1 1 1 1.04375 0 1 0 1 1 0 1 1 0 0 0 0 1 1 0 1.03750 0 1 0 1 1 1 0 0 0 0 0 1 0 0 1 Rev. 0 | Page 18 of 36
ADP3189
VR11 DAC CODES: VIDSEL = HIGH VR10.x DAC CODES: VIDSEL = LOW
OUTPUT VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6 1.03125 0 1 0 1 1 1 0 1 0 0 0 1 0 0 0 1.02500 0 1 0 1 1 1 1 0 0 0 0 1 0 1 1 1.01875 0 1 0 1 1 1 1 1 0 0 0 1 0 1 0 1.01250 0 1 1 0 0 0 0 0 0 0 0 1 1 0 1 1.00625 0 1 1 0 0 0 0 1 0 0 0 1 1 0 0 1.00000 0 1 1 0 0 0 1 0 0 0 0 1 1 1 1 0.99375 0 1 1 0 0 0 1 1 0 0 0 1 1 1 0 0.98750 0 1 1 0 0 1 0 0 0 0 1 0 0 0 1 0.98125 0 1 1 0 0 1 0 1 0 0 1 0 0 0 0 0.97500 0 1 1 0 0 1 1 0 0 0 1 0 0 1 1 0.96875 0 1 1 0 0 1 1 1 0 0 1 0 0 1 0 0.96250 0 1 1 0 1 0 0 0 0 0 1 0 1 0 1 0.95625 0 1 1 0 1 0 0 1 0 0 1 0 1 0 0 0.95000 0 1 1 0 1 0 1 0 0 0 1 0 1 1 1 0.94375 0 1 1 0 1 0 1 1 0 0 1 0 1 1 0 0.93750 0 1 1 0 1 1 0 0 0 0 1 1 0 0 1 0.93125 0 1 1 0 1 1 0 1 0 0 1 1 0 0 0 0.92500 0 1 1 0 1 1 1 0 0 0 1 1 0 1 1 0.91875 0 1 1 0 1 1 1 1 0 0 1 1 0 1 0 0.91250 0 1 1 1 0 0 0 0 0 0 1 1 1 0 1 0.90625 0 1 1 1 0 0 0 1 0 0 1 1 1 0 0 0.90000 0 1 1 1 0 0 1 0 0 0 1 1 1 1 1 0.89375 0 1 1 1 0 0 1 1 0 0 1 1 1 1 0 0.88750 0 1 1 1 0 1 0 0 0 1 0 0 0 0 1 0.88125 0 1 1 1 0 1 0 1 0 1 0 0 0 0 0 0.87500 0 1 1 1 0 1 1 0 0 1 0 0 0 1 1 0.86875 0 1 1 1 0 1 1 1 0 1 0 0 0 1 0 0.86250 0 1 1 1 1 0 0 0 0 1 0 0 1 0 1 0.85625 0 1 1 1 1 0 0 1 0 1 0 0 1 0 0 0.85000 0 1 1 1 1 0 1 0 0 1 0 0 1 1 1 0.84375 0 1 1 1 1 0 1 1 0 1 0 0 1 1 0 0.83750 0 1 1 1 1 1 0 0 0 1 0 1 0 0 1 0.83125 0 1 1 1 1 1 0 1 0 1 0 1 0 0 0 0.82500 0 1 1 1 1 1 1 0 N/A 0.81875 0 1 1 1 1 1 1 1 N/A 0.81250 1 0 0 0 0 0 0 0 N/A 0.80625 1 0 0 0 0 0 0 1 N/A 0.80000 1 0 0 0 0 0 1 0 N/A 0.79375 1 0 0 0 0 0 1 1 N/A 0.78750 1 0 0 0 0 1 0 0 N/A 0.78125 1 0 0 0 0 1 0 1 N/A 0.77500 1 0 0 0 0 1 1 0 N/A 0.76875 1 0 0 0 0 1 1 1 N/A 0.76250 1 0 0 0 1 0 0 0 N/A 0.75625 1 0 0 0 1 0 0 1 N/A 0.75000 1 0 0 0 1 0 1 0 N/A 0.74375 1 0 0 0 1 0 1 1 N/A 0.73750 1 0 0 0 1 1 0 0 N/A 0.73125 1 0 0 0 1 1 0 1 N/A
Rev. 0 | Page 19 of 36
ADP3189
VR11 DAC CODES: VIDSEL = HIGH VR10.x DAC CODES: VIDSEL = LOW
OUTPUT VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0 VID4 VID3 VID2 VID1 VID0 VID5 VID6 0.72500 1 0 0 0 1 1 1 0 N/A 0.71875 1 0 0 0 1 1 1 1 N/A 0.71250 1 0 0 1 0 0 0 0 N/A 0.70625 1 0 0 1 0 0 0 1 N/A 0.70000 1 0 0 1 0 0 1 0 N/A 0.69375 1 0 0 1 0 0 1 1 N/A 0.68750 1 0 0 1 0 1 0 0 N/A 0.68125 1 0 0 1 0 1 0 1 N/A 0.67500 1 0 0 1 0 1 1 0 N/A 0.66875 1 0 0 1 0 1 1 1 N/A 0.66250 1 0 0 1 1 0 0 0 N/A 0.65625 1 0 0 1 1 0 0 1 N/A 0.65000 1 0 0 1 1 0 1 0 N/A 0.64375 1 0 0 1 1 0 1 1 N/A 0.63750 1 0 0 1 1 1 0 0 N/A 0.63125 1 0 0 1 1 1 0 1 N/A 0.62500 1 0 0 1 1 1 1 0 N/A 0.61875 1 0 0 1 1 1 1 1 N/A 0.61250 1 0 1 0 0 0 0 0 N/A 0.60625 1 0 1 0 0 0 0 1 N/A 0.60000 1 0 1 0 0 0 1 0 N/A 0.59375 1 0 1 0 0 0 1 1 N/A 0.58750 1 0 1 0 0 1 0 0 N/A 0.58125 1 0 1 0 0 1 0 1 N/A 0.57500 1 0 1 0 0 1 1 0 N/A 0.56875 1 0 1 0 0 1 1 1 N/A 0.56250 1 0 1 0 1 0 0 0 N/A 0.55625 1 0 1 0 1 0 0 1 N/A 0.55000 1 0 1 0 1 0 1 0 N/A 0.54375 1 0 1 0 1 0 1 1 N/A 0.53750 1 0 1 0 1 1 0 0 N/A 0.53125 1 0 1 0 1 1 0 1 N/A 0.52500 1 0 1 0 1 1 1 0 N/A 0.51875 1 0 1 0 1 1 1 1 N/A 0.51250 1 0 1 1 0 0 0 0 N/A 0.50625 1 0 1 1 0 0 0 1 N/A 0.50000 1 0 1 1 0 0 1 0 N/A OFF 1 1 1 1 1 1 1 0 1 1 1 1 1 1 0 OFF 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1
Rev. 0 | Page 20 of 36
L1370nH18AVIN12V++VINRTNC1D21N41481232700μF / 16V / 3.3 A×2SANYO MV-WX SERIESR42.2ΩC918nFC2BSTDRVH8SW7PGND6Q1NTD40N03INODU2ADP3120C1110nFQ2NTD40N03560μF / 4V / 4V×10L2320nH/1.4mΩSANYO SEPC SERIES5mΩ EACHC124.7μF10Ω**C25++VCC(CORE)0.5V–1.6V115A TDC, 130A PKVCC(CORE) RTNC3410μF×18MLCCIN SOCKETVCC(SENSE)4VCC Q4NTD110N02Q3NTD110N02DRVL5C104.7μFR52.2ΩD11N4148C1318nFR110ΩD31N4148123U3ADP3120C1510nFBSTDRVH8SW7PGND6DRVL5INODVCCC164.7μFQ6NTD40N03L3320nH/1.4mΩQ5NTD40N0310Ω**VSS(SENSE)C3100μF(C3 OPTIONAL)C41μF4C144.7μFR62.2ΩC1718nFQ7NTD110N02 Q8NTD110N02VTT I/O40VIDSELVID0VID1VID2VID3VID4VID5VID6VID7VCC* FOR A DESCRIPTION OF OPTIONAL RSWRESISTORS, SEE THE THEORY OF OPERATION SECTION1C51nFU4ADP3120D41N4148123C1910nFDRVH8C204.7μFQ10NTD40N03L4320nH/1.4mΩPOWER GOODVRFANPROCHOTBSTINOD4SW7PGND6VCCDRVL500Figure 11.Typical 4-Phase Application Circuit
U1ADP3189FROM CPUCDLY18nFRSW4*RPH2RPH4158kΩ158kΩ1%1%CCS11nF5% NPOCCS2RCS1RCS2RPH31nF35.7kΩ88.7kΩ158kΩ5% NPO1%RPH1158kΩ1%RLIM100kΩ1%RTH1100kΩ, 5%NTCILIMITRTRAMPADJLLSETCSREFCSSUMCSCOMPGNDODNCRev. 0 | Page 21 of 36
RSW1*RSW2*RSW3*C184.7μFR72.2ΩC2118nFENPWRGDFBRTNFBCOMPSSDELAYVRFANVRHOTTTSENSEPWM1PWM2PWM3PWM4PWM5SW1SW2SW3SW4SW5CB560pFCFB27pFQ9NTD40N0310Ω**RB1.00kΩC60.1μFRACA560nF10.0kΩCSS39nF Q12NTD110N02Q11NTD110N02U5ADP3120D51N414812C2310nFBSTIN34C244.7μFDRVH8SW7ODVCCPGND6DRVL5RT182kΩ1%C71nFQ13NTD40N03Q14NTD40N03L5320nH/1.4mΩ10Ω**C224.7μFQ15NTD110N02 Q16NTD110N02RTH2100kΩ, 5%NTCC81nFR31Ω**CONNECT NEAREACHINDUCTORADP3189
ADP3189
APPLICATION INFORMATION
The design parameters for a typical Intel VRD 11 compliant CPU application are as follows: • • • • • •
Input voltage (VIN) = 12 V
VID setting voltage (VVID) = 1.300 V Duty cycle (D) = 0.108
Nominal output voltage at no load (VONL) = 1.285 V Nominal output voltage at 115 A load (VOFL) = 1.170 V Static output voltage drop based on a 1.0 mΩ load line (RO) from no load to full load (VD) = VONL − VOFL = 1.285 V − 1.170 V = 115 mV Maximum output current (IO) = 130 A Maximum output current step (ΔIO) = 100 A Maximum output current slew-rate (SR) = 200 A/μ sec Number of phases (n) = 4
Switching frequency per phase (fSW) = 330 kHz
SOFT START DELAY TIME
The value of CSS sets the soft start time. The ramp is generated with a 15 μA internal current source. The value for CSS can be found using:
CSS=15μA×
TD2
(2) VBOOT
• • • • •
where TD2 is the desired soft start time and VBOOT is internally set to 1.1 V. Assuming a desired TD2 time of 3 ms, CSS is 41 nF. The closest standard value for CSS is 39 nF. Although CSS also controls the time delay for TD4 (which is determined by the final VID voltage), the minimum specification for TD4 is 0 ns. This means that as long as the TD2 time requirement is met, TD4 will be within the specification.
CURRENT LIMIT LATCH-OFF DELAY TIMES
The start-up and current limit delay times are determined by the capacitor connected to the DELAY pin. The first step is to set CDLY for the TD1, TD3, and TD5 delay times (see Figure 8). The DELAY ramp (IDELAY) is generated using a 15 μA internal current source. The value for CDLY can be approximated using:
SETTING THE CLOCK FREQUENCY
The ADP3189 uses a fixed-frequency control architecture. The frequency is set by an external timing resistor (RT). The clock frequency and the number of phases determine the switching frequency per phase, which relates directly to switching losses and the sizes of the inductors, and of the input and output capacitors. With n = 4 for four phases, a clock frequency of 1.32 MHz sets the switching frequency (fSW) of each phase to 330 kHz, which represents a practical trade-off between the switching losses and the sizes of the out-put filter components. Equation 1 shows that to achieve a 1.32 MHz oscillator frequency, the correct value for RT is 181 kΩ. Alternatively, the value for RT can be calculated using
CDLY=IDELAY×
TD(x)VDELAY(TH)
(3)
where TD(x) is the desired delay time for TD1, TD3, and TD5. The DELAY threshold voltage (VDELAY(TH)) is given as 1.7 V. In this example, 2 ms is chosen for all three delay times, which meets Intel’s specification. Solving for CDLY gives a value of 17.6 nF. The closest standard value for CDLY is 18 nF.
When the ADP3189 goes into current limit, the internal current source changes from 15 μA to 3.75 μA. This makes the latch-off delay time 4 times longer than the start-up delay time. Longer latch-off delay times can be achieved by placing a resistor in 1
RT=−13kΩ (1)
parallel with CDLY. n×fSW×3.9pF
where 3.9 pF and 13 kΩ are internal IC component values.
For good initial accuracy and frequency stability, a 1% resistor is recommended.
Rev. 0 | Page 22 of 36
ADP3189
DESIGNING AN INDUCTOR
Once the inductance and DCR are known, the next step is to either design an inductor or to find a standard inductor that comes as close as possible to meeting the overall design goals. It is also important to have the inductance and DCR tolerance specified to control the accuracy of the system. 15% inductance and 7% DCR, at room temperature, are reasonable tolerances most manufacturers can meet.
The first decision in designing the inductor is choosing the core material. Several possibilities for providing low core loss at high frequencies include the powder cores (for example, Kool-Mμ® from Magnetics, Inc. or from Micrometals) and the gapped soft ferrite cores (for example, 3F3 or 3F4 from Philips). Low frequency powdered iron cores should be avoided due to their high core loss, especially when the inductor value is relatively low and the ripple current is high.
INDUCTOR SELECTION
The choice of inductance for the inductor determines the ripple current in the inductor. Less inductance leads to more ripple current, which increases the output ripple voltage and conduction losses in the MOSFETs, but allows using smaller inductors and, for a specified peak-to-peak transient deviation, less total output capacitance. Conversely, a higher inductance means lower ripple current and reduced conduction losses, but requires larger inductors and more output capacitance for the same peak-to-peak transient deviation.
In any multiphase converter, a practical value for the peak-to-peak inductor ripple current is less than 50% of the maximum dc current in the same inductor. Equation 4 shows the relationship between the inductance, oscillator frequency, and peak-to-peak ripple current in the inductor. Equation 5 can be used to determine the minimum inductance based on a given output ripple voltage.
The best choice for a core geometry is a closed-loop type such as a potentiometer core; PQ, U, or E core; or toroid. A good VVID×(1−D) (4) IR=compromise between price and performance is a core with
fSW×L
a toroidal shape.
VVID×RO×(1−(n×D))Many useful magnetics design references are available for (5) L≥
fSW×VRIPPLE
quickly designing a power inductor, such as
• •
Magnetic Designer Software Intusoft (www.intusoft.com)
Designing Magnetic Components for High-Frequency DC-DCConverters, by William T. McLyman, Kg Magnetics, Inc., ISBN 1883107008
Solving Equation 5 for an 8 mV p-p output ripple voltage yields
L≥
1.3V×1.0mΩ×(1−0.432)330kHz×8mV
=280nH
If the resulting ripple voltage is less than that designed for, the inductor can be made smaller until the ripple value is met. This allows optimal transient response and minimum output decoupling.
The smallest possible inductor should be used to minimize the number of output capacitors. For this example, choosing a 320 nH inductor is a good starting point and gives a calculated ripple current of 11 A. The inductor should not saturate at the peak current of 35.5 A and should be able to handle the sum of the power dissipation caused by the average current of 30 A in the winding and core loss.
Another important factor in the inductor design is the DCR (RL), which is used for measuring the phase currents. A large DCR can cause excessive power losses, while too small a value can lead to increased measurement error. A good rule is to have the DCR be about 1 to 1½ times the droop resistance (RO). This example uses an inductor with a DCR of 1.4 mΩ.
Selecting a Standard Inductor
The following power inductor manufacturers can provide design consultation and deliver power inductors optimized for high power applications upon request. • • • •
Coilcraft
www.coilcraft.com Coiltronics
www.coiltronics.com Sumida Electric Company www.sumida.com Vishay Intertechnology www.vishay.com
Rev. 0 | Page 23 of 36
ADP3189
CURRENT SENSE AMPLIFIER
Most designs require the regulator output voltage, measured at the CPU pins, to drop when the output current increases. The specified voltage drop corresponds to a dc output resistance (RO), also referred to as a load line. The ADP3189 has the flexibility of adjusting RO, independent of current limit or compensation components, and it can also support CPUs that do not require a load line.
For designs requiring a load line, the impedance gain of the CS amplifier (RCSA) must be to be greater than or equal to the load line. All designs, whether they have a load line or not, should keep RCSA ≥ 1 mΩ.
The output current is measured by summing the voltage across each inductor and passing the signal through a low-pass filter. This summer filter is the CS amplifier configured with resistors RPH(X) (summers), and RCS and CCS (filter). The impedance gain of the regulator is set by the following equations, where RL is the DCR of the output inductors:
RCSA=CCS=
INDUCTOR DCR TEMPERATURE CORRECTION
With the inductor’s DCR is used as the sense element and copper wire is the source of the DCR, the user needs to
compensate for temperature changes of the inductor’s winding. Fortunately, copper has a well known temperature coefficient (TC) of 0.39%/°C.
If RCS is designed to have an opposite and equal percentage change in resistance to that of the wire, it cancels the tempera-ture variation of the inductor’s DCR. Due to the nonlinear nature of NTC thermistors, resistors RCS1 and RCS2 are needed. See Figure 12 to linearize the NTC and produce the desired temperature tracking.
PLACE AS CLOSE AS POSSIBLETO THE NEAREST INDUCTORRTMRPH1RPH2RPH3ADP3189CSCOMP18RCS1CCS2RCS2KEEP THIS PATHAS SHORT ASPOSSIBLE ANDWELL AWAY FROMSWITCH NODE LINES05626-012RCSRPH(x)CCS1×RL (6) CSSUM17LCSREF (7) 16RL×RCS
The user has the flexibility of choosing either RCS or RPH(X). However, it is best to select RCS equal to 100 kΩ, and then solve for RPH(X) by rearranging Equation 6. Here RCSA = RO = 1 mΩ since this is equal to our design loadline.
RPH(x)=
RL
×RCS
RCSA
Figure 12. Temperature Compensation Circuit Values
The following procedure and expressions yield values to use for RCS1, RCS2, and RTH (the thermistor value at 25°C) for a given RCS value.
1. Select an NTC based on type and value. Since the value
is unknown, use a thermistor with a value close to RCS. The NTC should also have an initial tolerance of better than 5%. 2. Based on the type of NTC, find its relative resistance value
at two temperatures. The temperatures that work well are 50°C and 90°C. These resistance values are called
A (RTH(50°C))/RTH(25°C)) and B (RTH(90°C))/RTH(25°C)). The NTC’s relative value is always 1 at 25°C. 3. Find the relative value of RCS required for each of these
temperatures. This is based on the percentage change needed, which in this example is initially 0.39%/°C. These are called r1 (1/(1 + TC × (T1 − 25))) and r2 (1/(1 + TC × (T2 − 25))), where TC = 0.0039 for copper. T1 = 50°C and T2 = 90°C are chosen. From this, calculate that r1 = 0.9112 and r2 = 0.7978.
RPH(x)=
1.4mΩ1.0mΩ
×100kΩ=140kΩ
Next, use Equation 7 to solve for CCS.
CCS=
320nH1.4mΩ×100kΩ
=2.28nF
It is best to have a dual location for CCS in the layout so that standard values can be used in parallel to get as close to the value desired. For best accuracy, CCS should be a 5% or 10% NPO capacitor. This example uses a 5% combination for CCS of two 1 nF capacitors in parallel. Recalculating RCS and RPH(X) using this capacitor combination yields 114 kΩ and 160 kΩ. The closest standard 1% value for RPH(X) is 158 kΩ.
Rev. 0 | Page 24 of 36
ADP3189
LOAD LINE SETTING
For load line values greater than 1 mΩ, RCSA can be set equal (A−B)×r1×r2−A×(1−B)×r2+B×(1−A)×r1 (8)
to RO, and the LLSET pin can be directly connected to the A×(1−B)×r1−B×(1−A)×r2−(A−B)CSCOMP pin. When the load line value needs to be less than
(1−A)1 mΩ, two additional resistors are required. Figure 13 shows = (9)
A1−
1−rCS2r1−rCS2
111−
1−rCS2rCS1
1. Compute the relative values for RCS1, RCS2, and RTH using
rCS2=
rCS1
the placement of these resistors.
rTH=
(10) ADP3189CSCOMPCalculate RTH = rTH × RCS, then select the closest value of 17thermistor available. Also, compute a scaling factor k based
CSSUMon the ratio of the actual thermistor value used relative to 16the computed one:
CSREFRTH(ACTUAL)15k= (11) RTH(CALCULATED)RLL1RLL2OPTIONAL LOAD LINESELECT SWITCHRCS1=RCS×k×rCS1 (12) RCS2=RCS×((1−k)+(k×rCS2)) (13)
QLL05626-0132. Calculate values for RCS1 and RCS2 using Equation 12 and
Equation 13:
LLSET14 Figure 13. Load Line Setting Resistors
In this example, RCS was calculated to be 114 kΩ. Look for an available 100 kΩ thermistor, 0603 size. One such thermistor is the Vishay NTHS0603N01N1003JR NTC thermistor with A = 0.3602 and B = 0.09174. From these values, compute rCS1 = 0.3795, rCS2 = 0.7195, and rTH = 1.075.
Solving for RTH yields 122.55 kΩ, so 100 kΩ is chosen, making k = 0.816. Next find RCS1 and RCS2 to be 35.3 kΩ and 87.9 kΩ. Finally, choose the closest 1% resistor values, which yields a choice of 35.7 kΩ and 88.7 kΩ.
The two resistors RLL1 and RLL2 set up a divider between the
CSCOMP pin and CSREF pin. This resistor divider is input into the LLSET pin to set the load line slope RO of the VR according to the following equation:
RO=
RLL2
×RCSA (14)
RLL1+RLL2
For best results, start with a 1% resistor of 20.0 kΩ for RLL2. Then, solve for the required value of RLL1 by rearranging Equation 14 as follows:
RO=
RLL2
×RCSA
RLL1+RLL2
Another useful feature for some VR applications is the ability to select different load lines. Figure 13 shows an optional MOSFET switch that allows this. Here, design for RCSA = RO(MAX) (selected with QLL on) and then use Equation 14 to set RO = RO(MIN) (selected with QLL off).
For this design, RCSA = RO = 1 mΩ, so connect LLSET directly to CSCOMP, and the resistors RLL1 and RLL2 are not needed.
Rev. 0 | Page 25 of 36
ADP3189
OUTPUT OFFSET
The Intel specification requires that at no load the nominal output voltage of the regulator is offset to a value lower than the nominal voltage corresponding to the VID code. The offset is set by a constant current source flowing out of the FB pin (IFB) and flowing through RB. The value of RB can be found using Equation 15:
A lower limit is based on meeting the capacitance for load release for a given maximum load step ΔIO and a maximum allowable overshoot. The total amount of load release voltage is given as ΔVO = ΔIO × RO + ΔVrl, where ΔVrl is the maximum allowable overshoot voltage.
RB=
VVID−VONL
IFB
RB=
1.3V−1.285V
15μA
⎛⎞⎜⎟⎜⎟L×ΔIO
Cx(MIN)≥⎜−Cz⎟ (17)
ΔVrl⎞⎜n⎛⎟⎜RO+⎟×VVID×⎜⎟⎜⎟ΔIO⎠⎝⎝⎠
=1.00kΩ (15) Cx(MAX)≤ (18)
⎛
⎛VVIDnKROVVL⎜
⎜tv××1+⎜22⎜V×LnKROVVID⎜V⎝⎝⎛VERR
whereK=−1n⎜⎜V
⎝V
⎞⎟ ⎟⎠
The closest standard 1% resistor value is 1.00 kΩ.
COUT SELECTION
The required output decoupling for the regulator is typically recommended by Intel for various processors and platforms. Use some simple design guidelines to determine the require-ments. These guidelines are based on having both bulk capacitors and ceramic capacitors in the system.
First, select the total amount of ceramic capacitance. This is based on the number and type of capacitor to be used. The best location for ceramic capacitors is inside the socket, with 12 to 18 of size, 1206 being the physical limit. Other capacitors can be placed along the outer edge of the socket as well.
To aid in determining the minimum amount of ceramic capacitance required, start with a worst-case load step occur-ring right after a switching cycle has stopped. The ceramic capacitance then delivers the charge to the load while the load is ramping up and until the VR has responded with the next switching cycle.
The following equation gives the designer a rough approximation for determining the minimum ceramic capacitance needed. Due to the complexity of the PCB
parasitics and bulk capacitors, the actual amount of ceramic capacitance required may vary.
⎞⎟⎟⎠
2
⎞⎟
−1⎟−Cz ⎟⎠
To meet the conditions of these expressions and transient response, the ESR of the bulk capacitor bank (RX) should be less than two times the droop resistance (RO). If the CX(MIN) is larger than CX(MAX), the system cannot meet the VID on-the-fly speci-fication and can require the use of a smaller inductor or more phases (and may have to increase the switching frequency to keep the output ripple the same).
This example uses eighteen 10 μF 1206 MLC capacitors (CZ = 180 F). The VID on-the-fly step change is 450 mV in 230 μs with a setting error of 2.5 mV. The maximum allowable load release overshoot for this example is 50 mV, therefore solving for the bulk capacitance yields
⎛⎞⎜⎟⎜⎟320nH×100A
−180μF⎟=3.92mF≤⎜
⎜⎟⎛50mV⎞
⎟⎜×41.0mΩ1.3V×+⎜⎟⎟⎜⎜⎟100A⎠⎝⎝⎠
Cx(MIN)
320nH×450mV
C≤× ⎤⎡11⎛1⎞ΔIO (16) x(MAX)22
()4×5.2×1.0mΩ×1.3VCZ(MIN)≥×⎢×⎜−D⎟−⎥RO⎣fSW⎝n⎠2SR⎦
2⎛⎞
⎛⎞230μs1345210mΩ×.V××.×.⎜⎟The typical ceramic capacitors used are made up of multiple ⎟−1⎟−180μF⎜1+⎜⎜⎟450mV×320nH10 μF or 22 μF capacitors. For this example, Equation 16 yields ⎜⎟⎝⎠⎝⎠180.8 μF, so eighteen 10 uF ceramics will suffice.
Next, there is an upper limit imposed on the total amount of
bulk capacitance (CX) when the user considers the VID on-the-fly voltage stepping of the output (voltage step VV in time tV with error of VERR).
=43.0mFwhere K = 5.2
Rev. 0 | Page 26 of 36
ADP3189
Another important factor for the synchronous MOSFET is the input capacitance and feedback capacitance. The ratio of the feedback to input needs to be small (less than 10% is recom-mended) to prevent accidental turn-on of the synchronous MOSFETs when the switch node goes high.
Using ten 560 μF Al-Poly capacitors with a typical ESR of 6 mΩ each yields CX = 5.6 mF with an RX = 0.6 mΩ.
One last check should be made to ensure that the ESL of the bulk capacitors (LX) is low enough to limit the high frequency ringing during a load change.
Also, the time to switch the synchronous MOSFETs off should
This is tested using not exceed the nonoverlap dead time of the MOSFET driver
(40 ns typical for the ADP3120). The output impedance of the
Lx≤Cz×RO2×Q2
driver is approximately 2 Ω, and the typical MOSFET input gate
(19) 42resistances are about 1 Ω to 2 Ω, so a total gate capacitance of Lx≤180μF×(1mΩ)×=240pH
3less than 6000 pF should be adhered to. Since there are two
24
MOSFETs in parallel, the input capacitance for each synchronous where Q is limited to /3 to ensure a critically damped system.
MOSFET should be limited to 3000 pF. In this example, Lis approximately 240 pH for the ten
X
A1-Polys capacitors, which satisfies this limitation. If the LX
of the chosen bulk capacitor bank is too large, the number of ceramic capacitors may need to be increased, or lower ESL bulks used if there is excessive undershoot during a load transient. For this multimode control technique, all ceramic designs can be used providing the conditions of Equation 16, Equation 17, Equation 18, and Equation 19 are satisfied.
POWER MOSFETS
For this example, the N-channel power MOSFETs have been selected for one high-side switch and two low-side switches per phase. The main selection parameters for the power MOSFETs are VGS(TH), QG, CISS, CRSS, and RDS(ON). The minimum gate drive voltage (the supply voltage to the ADP3120) dictates whether standard threshold or logic-level threshold MOSFETs must be used. With VGATE ~10 V, logic-level threshold MOSFETs (VGS(TH) < 2.5 V) are recommended.
The maximum output current (IO) determines the RDS(ON) requirement for the low-side (synchronous) MOSFETs. With the ADP3189, currents are balanced between phases, thus the current in each low-side MOSFET is the output current divided by the total number of MOSFETs (nSF). With conduction losses being dominant, the following expression shows the total power being dissipated in each synchronous MOSFET in terms of the ripple current per phase (IR) and average total output current (IO):
The high-side (main) MOSFET has to be able to handle two main power dissipation components: conduction and switching losses. The switching loss is related to the amount of time it takes for the main MOSFET to turn on and off, and to the
current and voltage that are being switched. Basing the switching speed on the rise and fall time of the gate driver impedance and MOSFET input capacitance, the following expression provides an approximate value for the switching loss per main MOSFET, where nMF is the total number of main MOSFETs:
PS(MF)=2×fSW×
VCC×IO
nMF
×RG×
nMF
×CISS (21) n
where RG is the total gate resistance (2 Ω for the ADP3120 and about 1 Ω for typical high speed switching MOSFETs, making RG = 3 Ω), and CISS is the input capacitance of the main MOSFET. Adding more main MOSFETs (nMF) does not help the switching loss per MOSFET, since the additional gate capacitance slows switching. Use lower gate capacitance devices to reduce switching loss.
The conduction loss of the main MOSFET is given by the following, where RDS(MF) is the on resistance of the MOSFET:
PC(MF)⎡⎛I
O
=D×⎢⎜⎜n⎢⎣⎝MF
⎞1⎛n×IR
⎟+×⎜⎟⎜n12⎠⎝MF
2
⎞
⎟⎟⎠
2
⎤
⎥×RDS(MF) (22) ⎥⎦
PSF
⎡⎛I
O
=(1−D)×⎢⎜⎜n⎢⎣⎝SF
⎞1⎛nIR
⎟+×⎜⎟12⎜⎠⎝nSF
2
⎞
⎟⎟⎠
2
⎤
⎥×RDS(SF) (20) ⎥⎦
Knowing the maximum output current being designed for and
the maximum allowed power dissipation, the user can find the required RDS(ON) for the MOSFET. For D-PAK MOSFETs up to an ambient temperature of 50°C, a safe limit for PSF is 1 W to 1.5 W at 120°C junction temperature. Thus, for this example (119 A maximum), RDS(SF) (per MOSFET) < 7.5 mΩ. This RDS(SF) is also at a junction temperature of about 120°C, so be certain to account for this when making this selection. This example uses two lower-side MOSFETs at 4.8 mΩ, each at 120°C.
Typically, for main MOSFETs, the highest speed (low CISS) device is preferred, but these usually have higher on resistance. Select a device that meets the total power dissipation (about 1.5 W for a single D-PAK) when combining the switching and conduction losses.
Rev. 0 | Page 27 of 36
ADP3189
For this example, an NTD40N03L was selected as the main MOSFET (eight total; nMF = 8), with CISS = 584 pF (max) and RDS(MF) = 19 mΩ (max at TJ = 120°C), and an NTD110N02L was selected as the synchronous MOSFET (eight total; nSF = 8), with CISS = 2710 pF (max) and RDS(SF) = 4.8 mΩ (max at TJ = 120°C). The synchronous MOSFET CISS is less than 3000 pF, satisfying this requirement. Solving for the power dissipation per MOSFET at IO = 119 A and IR = 11 A yields 958 mW for each synchronous MOSFET and 872 mW for each main MOSFET. The guideline is to limit the MOSFET power dissipation to 1 W. The values calculated in Equation 21 and Equation 22 comply with this guideline.
Finally, consider the power dissipation in the driver for each phase. This is best expressed as QG for the MOSFETs and is given by the following equation, where QGMF is the total gate charge for each main MOSFET and QGSF is the total gate charge for each synchronous MOSFET.
The internal ramp voltage magnitude can be calculated by using
VR=
AR×(1−D)×VVIDRR×CR×fSW
(25)
0.2×(1−0.108)×1.3V357kΩ×5pF×330kHz
VR=
=394mV
The size of the internal ramp can be made larger or smaller. If it is made larger, stability and noise rejection improves, but transient degrades. Likewise, if the ramp is made smaller, transient response improves at the sacrifice of noise rejection and stability.
The factor of 3 in the denominator of Equation 24 sets a ramp size that gives an optimal balance for good stability, transient response, and thermal balance.
⎡f⎤
PDRV=⎢SW×(nMF×QGMF+nSF×QGSF)+ICC⎥×VCC(23)
⎢⎥⎣2×n⎦Also shown is the driver’s standby dissipation factor (ICC × VCC).
For the ADP3120, the maximum dissipation should be less than 400 mW. In this example, with ICC = 7 mA, QGMF = 5.8 nC, and QGSF = 48 nC, one finds 297 mW in each driver, which is below the 400 mW dissipation limit. See the ADP3120 data sheet for more details.
COMP PIN RAMP
A ramp signal on the COMP pin is due to the droop voltage and output voltage ramps. This ramp amplitude adds to the internal ramp to produce the following overall ramp signal at the PWM input:
VRT=
⎛2×(1−n×D)⎜1−
⎜n×f×C×R
XSWO⎝
VR
⎞
⎟⎟⎠
(26)
RAMP RESISTOR SELECTION
The ramp resistor (RR) is used for setting the size of the internal PWM ramp. The value of this resistor is chosen to provide the best combination of thermal balance, stability, and transient response. The following expression is used for determining the optimum value:
RR=
AR×L3×AD×RDS×CR
In this example, the overall ramp signal is 0.46 V. However, if the ramp size is smaller than 0.5 V, increase the ramp size to be at least 0.5 V by decreasing the ramp resistor for noise immunity. As there is only 0.46 V initially, a ramp resistor value of 332 kΩ is chosen for this example, yielding an overall ramp of 0.51 V.
(24)
RR=where
AR is the internal ramp amplifier gain. AD is the current balancing amplifier gain.
RDS is the total low-side MOSFET on resistance. CR is the internal ramp capacitor value.
0.2×320nH3×5×2.4mΩ×5pF
=356kΩ
Rev. 0 | Page 28 of 36
ADP3189
FEEDBACK LOOP COMPENSATION DESIGN
CURRENT LIMIT SETPOINT
To select the current limit setpoint, first find the resistor value for RLIM. The current limit threshold for the ADP3189 is set with a 1.7 V source (VLIM) across RLIM with a gain of 10 mV/μA (ALIM). RLIM can be found using
Optimized compensation of the ADP3189 allows the best possible response of the regulator’s output to a load change. The basis for determining the optimum compensation is to make the regulator and output decoupling appear as an output impedance that is entirely resistive over the widest possible ALIM×VLIM
RLIM= (27) frequency range, including dc, and equal to the droop resis-ILIM×RCSA
tance (RO). With the resistive output impedance, the output voltage droops in proportion to the load current at any load For values of RLIM greater than 500 kΩ, the current limit may be
current slew rate. This ensures the optimal positioning and lower than expected, so some adjustment of RLIM is needed. Here,
allows the minimization of the output decoupling. I is the peak average current limit for the supply output. In this
LIM
example, choosing a peak current limit of 170 A for ILIM, results
in RLIM = 100 kΩ, and 100 kΩ is chosen as the nearest 1% value.
With the multimode feedback structure of the ADP3189, the feedback compensation must be set to make the converter’s output impedance, working in parallel with the output decoup-The per-phase initial duty cycle limit and peak current during a
ling, to meet this goal. Several poles and zeros created by the load step are determined by
output inductor and decoupling capacitors (output filter) need VCOMP(MAX)−VBIAS
DMAX=D× (28) to be compensated for.
VRT
IPHMAX≅
A type-three compensator on the voltage feedback is adequate DMAX(VIN−VVID) (29)
×for proper compensation of the output filter. Equation 31 to fSWL
Equation 35 are intended to yield an optimal starting point for
the design; some adjustments may be necessary to account for PCB and component parasitic effects (see the Tuning the ADP3189 section).
For the ADP3189, the maximum COMP voltage (VCOMP(MAX)) is 4.0 V and the COMP pin bias voltage (VBIAS) is 1.1 V. In this example, the maximum duty cycle is 0.61 and the peak current is 62 A.
The limit of the peak per-phase current described earlier during the secondary current limit is determined by
IPHLIM
First, compute the time constants for all the poles and zeros in the system, using Equation 31 to Equation 35 on the next page.
VCOMP(CLAMPED)−VBIAS (30) ≅
AD×RDS(MAX)For the ADP3189, the current balancing amplifier gain (AD) is 5, and the clamped COMP pin voltage is 2 V. Using an RDS(MAX) of 2.8 mΩ (low-side on resistance at 150°C) results in a per-phase peak current limit of 64 A. This current level can be reached only with an absolute short at the output, and the current limit latch-off function shuts down the regulator before overheating can occur.
Rev. 0 | Page 29 of 36
ADP3189
The first step is to compute the time constants for all of the poles and zeros in the system:
RE=n×RO+AD×RDS+
RL×VRTVVID
+
2×L×(1−n×D)×VRTn×CX×RO×VVID
+
RE=4×1mΩ+5×2.4mΩ+
1.4mΩ×0.51V
1.3V
2×320nH×(1−0.432)×0.51V4×5.6mF×1mΩ×1.3V
=22.9mΩ (31)
TA=CX×(RO−R')+
240pH1mΩ−0.5mΩLXRO−R'
=3.00μs (32) ×=5.6mF×(1mΩ−0.5mΩ)+×
1mΩ0.6mΩRORX
TB=(RX+R'−RO)×CX=(0.6mΩ+0.5mΩ−1mΩ)×5.6mF=560ns (33) ⎛AD×RDS
VRT×⎜L−
⎜2×fSW⎝
TC=
VVID×RETD=
2
CX×CZ×RO
⎞⎟⎟⎠
⎛5×2.4mΩ⎞
⎟0.51V×⎜320nH−
⎜⎟×2330kHz⎝⎠
==5.17μs (34)
1.3V×22.9mΩ=
5.6mF×(1mΩ−0.5mΩ)+180μF×1mΩ
5.6mF×180μF×(1mΩ)2
CX×(RO−R')+CZ×RO
=338ns (35)
where, for the ADP3189, R' is the PCB resistance from the bulk capacitors to the ceramics and where RDS is the total low-side MOSFET on resistance per phase. In this example, AD is 5, VRT equals 0.51 V, R' is approximately 0.5 mΩ (assuming a 4-layer, 1 ounce mother-board), and LX is 240 pH for the ten Al-Poly capacitors. The compensation values can then be solved using
CA=RA=
n×RO×TARE×RB
=
4×1mΩ×3.00μs22.9mΩ×1.00kΩ
=524pF (36)
5.17μsTC==9.87kΩ CA524pF
(37) (38) (39)
CB=CFB=
TB560ns
==560pF RB1.00kΩ
TD338ns
==34.2pF RA9.87kΩ
These are the starting values prior to tuning the design to account for layout and other parasitic effects (see the Tuning the ADP3189 section). The final values selected after tuning are
CA = 560 pF RA = 10.0 kΩ CB = 560 pF CFB = 27 pF
Rev. 0 | Page 30 of 36
ADP3189
The capacitor manufacturer’s ripple current ratings are often based on only 2,000 hours of life. This makes it advisable to further derate the capacitor or to choose a capacitor rated at a higher temperature than required. Several capacitors may be placed in parallel to meet size or height requirements in the design. In this example, the input capacitor bank is formed by two 2,700 μF, 16 V aluminum electrolytic capacitors, and eight 4.7 μF ceramic capacitors.
To reduce the input current di/dt to a level below the recom-mended maximum of 0.1 A/μs, an additional small inductor (L > 370 nH at 18 A) should be inserted between the converter and the supply bus. This inductor also acts as a filter between the converter and the primary power source.
Figure 14 and Figure 15 show the typical transient response using these compensation values.
05626-014THERMAL MONITOR DESIGN
A thermistor is used on the TTSENSE input of the ADP3189 for monitoring the temperature of the VR. A constant current of 120 μA is sourced out of this pin and run through a thermis-tor network such as that shown in Figure 16.
ADP3189VRFAN8Figure 14. Typical Transient Response for Design Example
Load Step
VRHOT9TTSENSE10OPTIONALTEMPERATUREADJUSTRESISTORPLACE THERMISTORNEAR CLOSEST PHASE05626-0150.105626-016RTTSENSE
Figure 15. Typical Transient Response for Design Example Load Release
Figure 16. VR Thermal Monitor Circuit
CIN SELECTION AND INPUT CURRENT di/dt REDUCTION
In continuous inductor current mode, the source current of the high-side MOSFET is approximately a square wave with a duty ratio equal to n × VOUT/VIN and an amplitude of one-nth the maximum output current. To prevent large voltage transients, a low ESR input capacitor, sized for the maximum rms current, must be used. The maximum rms capacitor current is given by
A voltage is generated from this current through the thermistor and sensed inside the IC. When the voltage reaches 1.11 V, the VRFAN output gets set. When the voltage reaches 0.81 V, the VRHOT gets set. This corresponds to RTTSENSE values of 9.25 kΩ for VRFAN and 6.75 kΩ.
These values correspond to a thermistor temperature of ~100°C and ~110°C when using the same type of 100 kΩ NTC thermistor used in the current sense amplifier.
ICRMS=D×IO×
1−1N×D
1−1=14.7A
4×0.108
An additional fixed resistor in parallel with the thermistor provides tuning the trip point temperatures to match the hot-test temperature in the VR, when the thermistor itself is directly sensing a proportionately lower temperature. Setting this resistor value is best accomplished with a variable resistor
(40) during thermal validation, and then fixing this value for the
final design.
Additionally, a 0.1 μF should be used for filtering noise.
ICRMS=0.108×119A×
Rev. 0 | Page 31 of 36
ADP3189
TUNING THE ADP3189
1. Build a circuit based on the compensation values
computed from the design spreadsheet. 2. Hook up the dc load to circuit, turn it on, and verify its
operation. Also, check for jitter at no load and full load.
6. Measure the output voltage from no load to full load using
5 A steps. Compute the loadline slope for each change, and then average to get overall loadline slope (ROMEAS). 7. If ROMEAS is off from RO by more than 0.05 mΩ, use the
following to adjust the RPH values:
RPH(NEW)=RPH(OLD)×
DC Loadline Setting
3. Measure the output voltage at no load (VNL). Verify that it
is within tolerance. 4. Measure the output voltage at full load cold (VFLCOLD). Let
the board sit for ~10 minutes at full load, and then measure the output (VFLHOT). If there is a change of more than a few millivolts, adjust RCS1 and RCS2 using Equation 41 and Equation 43.
RCS2(NEW)=RCS2(OLD)×
ROMEAS
(42) RO
8. Repeat Step 6 and Step 7 to check the loadline, and repeat
adjustments if necessary. 9. Once dc loadline adjustment is complete, do not change
RPH, RCS1, RCS2, or RTH for the remainder of the procedure.
10. Measure the output ripple at no load and full load with VNL−VFLCOLD
(41) a scope, and make sure it is within specifications.
VNL−VFLHOT
5. Repeat Step 4 until the cold and hot voltage measurements
remain the same.
RCS1(NEW)=
1
RCS1(OLD)×RTH(25°C)+RCS1(OLD)−RCS2(NEW)×RCS1(OLD)−RTH(25°C)(RCS1(OLD)+RTH(25°C))()−R
1
TH(25°C) (43)
Rev. 0 | Page 32 of 36
ADP3189
Initial Transient Setting
18. With the dynamic load still set at the maximum step size,
expand the scope time scale to see 2 μs/div to 5 μs/div. The waveform can have two overshoots and one minor undershoot (see Figure 18). Here, VDROOP is the final desired value.
AC Loadline Setting
11. Remove the dc load from the circuit and hook up the
dynamic load. 12. Hook up the scope to the output voltage and set it to dc
coupling with the time scale at 100 μs/div. 13. Set the dynamic load for a transient step of about 40 A at
1 kHz with 50% duty cycle. 14. Measure the output waveform (use dc offset on scope to
see the waveform). Try to use a vertical scale of
100 mV/div or finer. This waveform should look similar to Figure 17.
VDROOPVTRAN1VTRAN205626-018VACDRPVDCDRP
Figure 18. Transient Setting Waveform
Figure 17. AC Loadline Waveform
19. If both overshoots are larger than desired, try making
the adjustments described later in this step. If these adjustments do not change the response, you are limited by the output decoupling. Check the output response each time you make a change, and check the switching nodes to make ensure that the response is still stable.
• • •
Make the ramp resistor larger by 25% (RRAMP). For VTRAN1, increase CB or increase the switching frequency.
For VTRAN2, increase RA and decrease CA by 25%.
15. Use the horizontal cursors to measure VACDRP and VDCDRP as
shown. Do not measure the undershoot or overshoot that happens immediately after this step. 16. If VACDRP and VDCDRP are different by more than a few
millivolts, use Equation 44 to adjust CCS. You may need to parallel different values to get the right one since there are limited standard capacitor values available. (It is a good idea to have locations for two capacitors in the layout for this.)
CCS(NEW)=CCS(OLD)×
VACDRPVDCDRP
20. For load release (see Figure 19), if VTRANREL is larger
than the allowed overshoot, there is not enough output capacitance. Either more capacitance is needed, or the inductor values need to be made smaller. (When changing inductors, start the design again using a spreadsheet and
(44) this tuning procedure.)
17. Repeat Step 11 to Step 13 and repeat the adjustments if
necessary. Once complete, do not change CCS for the remainder of the procedure.
Set the dynamic load step to maximum step size (do not use a step size larger than needed) and verify that the output waveform is square, which means that VACDRP and VDCDRP are equal.
05626-017VTRANRELVDROOP05626-019
Figure 19. Transient Setting Waveform
Rev. 0 | Page 33 of 36
ADP3189
Since the ADP3189 turns off all of the phases (switches inductors to ground), there is no ripple voltage present during load release. Therefore, the user does not have to add headroom for ripple, allowing load release VTRANREL to be larger than VTRAN1, by the amount of ripple, and still meet specifications.
If VTRAN1 and VTRANREL are less than the desired final droop, this implies that capacitors can be removed. When removing capaci-tors, check the output ripple voltage as well to make sure it is still within specifications.
Avoid crossing any signal lines over the switching power path loop, described in the Power Circuitry Recommendations section.
Power Circuitry Recommendations
The switching power path should be routed on the PCB to encompass the shortest-possible length in order to minimize radiated switching noise energy (that is, EMI) and conduction losses in the board. Failure to take proper precautions often results in EMI problems for the entire PC system and noise-related operational problems in the power converter control circuitry. The switching power path is the loop formed by the current path through the input capacitors and the power MOSFETs, including all interconnecting PCB traces and planes. Using short and wide interconnection traces is especially critical in this path for two reasons: it minimizes the inductance in the switching loop, which can cause high energy ringing, and it accommodates the high current demand with minimal voltage loss.
Whenever a power dissipating component, for example, a power MOSFET, is soldered to a PCB, the liberal use of vias, both directly on the mounting pad and immediately surrounding it, is recommended. Two important reasons for this are improved current rating through the vias and improved thermal perform-ance from vias extended to the opposite side of the PCB, where a plane can more readily transfer the heat to the air. Make a mirror image of any pad being used to heatsink the MOSFETs on the opposite side of the PCB to achieve the best thermal dissipation to the air around the board. To further improve thermal performance, use the largest possible pad area. The output power path should also be routed to encompass a short distance. The output power path is formed by the current path through the inductor, the output capacitors, and the load. For best EMI containment, a solid power ground plane should be used as one of the inner layers extending fully under all the power components.
LAYOUT AND COMPONENT PLACEMENT
The following guidelines are recommended for optimal performance of a switching regulator in a PC system.
General Recommendations
For good results, a PCB with at least four layers is recommended. This should allow the needed versatility for control circuitry interconnections with optimal placement, power planes for ground, input and output power, and wide interconnection traces in the remainder of the power delivery current paths. Keep in mind that each square unit of 1 ounce copper trace has a resistance of ~0.53 mΩ at room temperature.
Whenever high currents must be routed between PCB layers, vias should be used liberally to create several parallel current paths, so the resistance and inductance introduced by these current paths is minimized and the via current rating is not exceeded.
If critical signal lines (including the output voltage sense lines of the ADP3189) must cross through power circuitry, it is best if a signal ground plane can be interposed between those signal lines and the traces of the power circuitry. This serves as a shield to minimize noise injection into the signals at the expense of making signal ground a bit noisier.
An analog ground plane should be used around and under the ADP3189 as a reference for the components associated with the controller. This plane should be tied to the nearest output
decoupling capacitor ground and should not be tied to any other power circuitry to prevent power currents from flowing in it. The components around the ADP3189 should be located close to the controller with short traces. The most important traces to keep short and away from other traces are the FB pin and CSSUM pin. The output capacitors should be connected as close as possible to the load (or connector), for example, a microprocessor core, that receives the power. If the load is distributed, the capacitors should also be distributed and generally be in proportion to where the load tends to be more dynamic.
Signal Circuitry Recommendations
The output voltage is sensed and regulated between the FB pin and the FBRTN pin, which connect to the signal ground at the load. To avoid differential mode noise pickup in the sensed signal, the loop area should be small. Thus, the FB trace and FBRTN trace should be routed adjacent to each other on top of the power ground plane back to the controller.
The feedback traces from the switch nodes should be connected as close as possible to the inductor. The CSREF signal should be connected to the output voltage at the nearest inductor to the controller.
Rev. 0 | Page 34 of 36
ADP3189
OUTLINE DIMENSIONS
6.00BSC SQ0.60 MAX0.60 MAX3130401PIN 1INDICATORPIN 1INDICATORTOPVIEW5.75BCS SQ0.50BSC0.500.400.30EXPOSEDPAD(BOTTOM VIEW)4.254.10 SQ3.95101121200.25 MIN4.50REF12° MAX0.80 MAX0.65 TYP0.05 MAX0.02 NOM1.000.850.80SEATINGPLANE0.300.230.180.20 REFCOPLANARITY0.08COMPLIANT TO JEDEC STANDARDS MO-220-VJJD-2
Figure 20. 40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
6 mm × 6 mm Body, Very Thin Quad
(CP-40)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP3189JCPZ-RL1
Temperature Range 0°C to 85°C Package Description
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
Ordering Quantity Package Option
CP-40 2500
1
Z = Pb-free part.
Rev. 0 | Page 35 of 36
ADP3189
NOTES
© 2005 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05626–0–7/05(0)
Rev. 0 | Page 36 of 36
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